Wireless sensor reader

ABSTRACT

A wireless sensor reader is provided to interface with a wireless sensor. The wireless sensor reader transmits a narrowband, fixed frequency excitation pulse to cause the wireless sensor to generate a ring signal. The ring signal corresponds to the value of the physical parameter being sensed. The wireless sensor reader receives and amplifies the ring signal and sends the signal to a phase-locked loop. A voltage-controlled oscillator in the phase-locked loop locks onto the ring signal frequency and generates a count signal at a frequency related to the ring signal frequency. The voltage-controlled oscillator is placed into a hold mode where the control voltage is maintained constant to allow the count signal frequency to be determined. The low power, simple circuitry required to generate the excitation pulse allows the reader to be a small, battery operated unit. Alternative methods of frequency determination are also disclosed.

RELATED APPLICATIONS

This non-provisional application is a continuation of U.S. patentapplication Ser. No. 12/727,306 filed on Mar. 19, 2010 which is acontinuation-in-part of U.S. patent application Ser. No. 12/419,326filed on Apr. 7, 2009, now U.S. Pat. No. 8,154,389 which is acontinuation-in-part of U.S. patent application Ser. No. 12/075,858filed on Mar. 14, 2008, now abandoned which claims priority to U.S.Provisional Application No. 60/918,164 filed on Mar. 15, 2007, each ofwhich is hereby incorporated by reference.

TECHNICAL FIELD

This invention relates generally to reading passive wireless sensors,and more particularly to a reader circuitry and a method for excitingand sensing data from passive wireless sensors.

BACKGROUND

Passive wireless sensor systems that employ resonant circuit technologyare known. These systems utilize a passive wireless sensor in remotecommunication with excitation and reader circuitry. Often the wirelesssensor is implanted at a specific location, such as within the humanbody, to detect and report a sensed parameter. The sensed parametervaries the resonant circuit frequency of the wireless sensor. The readerdevice samples the resonant frequency of the wireless sensor todetermine the sensed parameter.

Early researcher Haynes (H. E. Haynes and A. L. Witchey, “Medicalelectronics, the pill that ‘talks’”, RCA Engineer, vol 5, pp. 52-54.1960) discloses an ingestible pill incorporating a wireless pressuresensor, with a large reader device surrounding the subject's body andmeasuring frequency by means of a discriminator circuit. Nagumo (J.Nagumo, A. Uchiyama, S. Kimoto, T. Watanuki, M. Hori, K. Suma, A. Ouchi,M. Kumano, and H. Watanabe, “Echo capsule for medical use (a batterylessradioendosonde)”, IRE Transactions on Bio-Medical Electronics. vol.BME-9, pp. 195-199, 1962) discloses a similar system, in which thesensor includes an energy storing capacitor to power the sensor duringresonance.

U.S. Pat. No. 4,127,110 by Bullara discloses a sensor for measuringbrain fluid pressure. U.S. Pat. No. 4,206,762 by Cosman discloses asimilar sensor for measuring intra-cranial pressure. Specifically, theCosman patent describes the use of a grid dip system for wirelesslymeasuring the resonant frequency of the sensor.

Several methods of reading passive wireless sensors have also beendescribed in prior patents. For example, the Cosman patent discloses anexternal oscillator circuit that uses the implanted sensor for tuning,and a grid dip measurement system for measurement of sensor resonantfrequency. U.S. Pat. No. 6,015,386 by Kensey, et al., discloses a readerthat excites the passive sensor by transmitting frequency sweeps anduses a phase detector on the transmit signal to identify the pointduring the sweep at which the transmitted frequency matches the resonantfrequency of the sensor. U.S. Pat. No. 6,206,835 by Spillman, et al.,discloses a medical implant application for reader technology disclosedin U.S. Pat. No. 5,581,248 by Spillman, et al. This reader technologydetects a frequency dependent variable impedance loading effect on thereader by the sensor's detected parameter. U.S. Pat. No. 7,432,723 byEllis, et al., discloses a reader with energizing loops each tuned toand transmitting different frequencies spaced to ensure that thebandwidth of the sensor allows resonant excitation of the sensor. Ellisuses a ring-down response from the appropriate energizing loop todetermine the sensor resonant frequency. U.S. Pat. No. 6,111,520 byAllen, et. al., discloses a method of transmitting a “chirp” of whitenoise to the sensor and detecting the ring-down response.

Some readers utilize phased-locked-loop (“PLL”) circuitry to lock ontothe sensor's resonant frequency. U.S. Pat. No. 7,245,117 by Joy, et al.discloses an active PLL circuit and signal processing circuit thatadjusts a transmitting PLL frequency until the received signal phase andthe transmitting PLL signal phase match. When this match occurs, thetransmitting PLL frequency is equal to the sensor resonant frequency.

PLL circuits may incorporate sample and hold (S/H) functions to samplethe input frequency and hold the PLL at a given frequency. PLLs with S/Hmay be used in a variety of applications. For example, U.S. Pat. No.4,531,526 by Genest discloses a reader that uses a PLL circuit with aS/H circuit to adjust the transmitted frequency of the reader to matchthe resonant frequency received from the sensor. This is done tomaximize sensor response to the next transmission and measures the decayrate of the sensor resonance amplitude to extract the sensed parametervalue. U.S. Pat. No. 4,644,420 by Buchan describes a PLL with a S/H usedto sample a tape data stream and maintain an appropriate samplingfrequency for evaluation of digital data pulses on the tape. U.S. Pat.No. 5,006,819 by Buchan, et al., provides additional enhancements tothis concept. U.S. Pat. No. 5,920,233 by Denny describes a high-speedsampling technique using a S/H circuit with a PLL to reduce the chargepump noise from the phase-frequency detector to enhance the low jitterperformance of a frequency synthesizing circuit. U.S. Pat. No. 4,511,858by Charavit, et al., discloses a PLL with a S/H circuit to pre-positionthe control voltage of a voltage controlled oscillator when the PLL lockfrequency is being changed. This is done to enhance the response speedof the PLL when changing the desired synthesized frequency. U.S. Pat.No. 6,570,457 by Fischer and U.S. Pat. No. 6,680,654 by Fischer, et al.,disclose a PLL with S/H circuitry to enhance PLL frequency stepping, aswell as an offset correction feature. U.S. Pat. No. 3,872,455 by Fuller,et al. discloses a PLL having a digital S/H to freeze the frequencydisplay and preload the frequency counter when a PLL phase lock isdetected.

Readers have also been found that implement direct signal sampling andfrequency analysis techniques. One example is U.S. Pat. No. 7,048,756 byEggers, et al., which measures internal body temperature using aresonant sensor with a curie temperature to show response change at atemperature threshold.

Further, readers using digital signal analysis to improve performanceand response are known. U.S. Pat. No. 7,466,120 by Miller, et al.,describes using a digital signal processor (DSP) to evaluate theresponse of a passive blood pressure sensor that has been excited by afrequency pulse then evaluating response signals from a triple-frequencyexcitation for relative phase delays.

Current designs for passive sensor readers, such as those disclosedabove, suffer from a number of deficiencies. The early “pulsed echoringing systems” of Haynes and Nagumo required large, high-poweredreader devices. Additionally, Collins (C. Collins, “Miniature PassivePressure Transensor for Implanting in the Eye”, IEEE Transactions onBio-Medical Engineering, vol BME-14, no. 2, April 1967) discloses thatthese systems suffered from inaccuracy and poor resolution due todifficulties in measuring the short-lived ring signal frequency, leadingto their abandonment in favor of various swept-frequency methods.

Swept frequency sensor readers similar to those described in the Cosman,Kensey, Ellis and Spillman patents, as well as the pulse methoddescribed by Allen, require relatively wide bandwidth allowance by thegovernment body regulating radio transmissions. This limits other usesof the spectrum and makes interference a potential issue. Readers thattrack the resonant frequency of a passive resonant sensor with avariable frequency transmitter, such as Genest, Ellis, and Joy alsosuffer from similar problems. The additional circuitry required byswept-frequency and/or digital tracking approaches is significant,adding to reader size, cost, and failure rate. Moreover, the amount ofelectrical power needed for transmissions, signal processing, sampling,and tracking the resonant frequency of a sensor using digitallycontrolled frequency tracking or swept frequency systems is significantand limits the ability to use battery power in a reader, as well aslimiting the longevity of batteries in a battery powered reader.Accordingly, an improved passive sensor and reader system is needed inthe art.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference to the detailed description is taken in connection with thefollowing illustrations:

FIG. 1 is a block diagram of a passive wireless sensor system;

FIG. 2 is a flow diagram illustrating the process of acquiring a readingfrom the sensor;

FIG. 3 is a plot qualitatively illustrating the frequencycharacteristics of the signal exchange between the sensor and thereader;

FIGS. 4a, 4b, and 4c illustrate three sequential plots qualitativelyeach showing the frequency characteristics of the signal exchangebetween sensor and reader during a reading acquisition;

FIG. 5 is a block diagram of the passive wireless sensor system of FIG.1, expanded to include an external data interface and remote datahandling functions;

FIG. 6 is a block diagram of the passive wireless sensor system of FIG.1, augmented by an intermediate antenna;

FIG. 7 is a top-level block diagram of the reader internal circuitry;

FIG. 8 is a block diagram of the timing and control portion of thereader circuitry;

FIG. 9 is a block diagram of the transmit portion of the readercircuitry;

FIG. 10 is a block diagram of the receive portion of the readercircuitry;

FIG. 11 is a block diagram of the phase locked loop portion of thereader circuitry;

FIG. 12 is a block diagram of the frequency counter portion of thereader circuitry;

FIG. 13 is a block diagram of an alternate embodiment of the phaselocked loop portion of the reader circuitry shown in FIG. 11, with adigital sampling timer and generation functionality for implementing thesample and hold;

FIG. 14 is a block diagram depicting an alternate embodiment of thereader internal circuitry of FIG. 7, in which the PLL and FrequencyCounter are replaced by Digital Sampling Circuitry and Spectrum AnalysisCircuitry;

FIG. 15 is a block diagram depicting an alternate embodiment of thetiming and control circuitry of FIG. 8, in which the PLL timer andFrequency Counter timer are replaced by a Digital Sampling timer andSpectrum Analysis timer, respectively;

FIG. 16 is a block diagram of the internal architecture of the DigitalSampling Circuitry block of FIG. 14; and

FIG. 17 is a block diagram of the internal architecture of the SpectrumAnalysis Circuitry block of FIG. 14.

SUMMARY

A reader device is provided to interface with a wireless sensor, whoseresonant frequency varies proportionately with the sensed parameter. Thereader transmits a short pulse of energy at a fixed frequency to causethe wireless sensor to ring at or near its resonant frequencyimmediately after the transmission ends. The reader receives andamplifies the sensor ring signal, and measures its frequency. In oneembodiment, the reader carries out this measurement by sending thesignal to a phase-locked loop (“PLL”) that locks to the sensor ringfrequency. Once the PLL has locked to the ring frequency, the PLL'svoltage controlled oscillator (“VCO”) is placed in a hold mode tomaintain the VCO frequency at the locked frequency. The VCO frequency iscounted to determine the sensor resonant frequency. Alternately, the VCOcontrol voltage itself is sampled and is used to determine sensorresonant frequency based on a known correlation. When VCO controlvoltage is sampled, the VCO frequency may not need to be locked if thevoltage sampling is sufficiently fast. Further frequency determinationmethods and systems involving digital spectrum analysis are alsodisclosed.

DETAILED DESCRIPTION

A passive wireless sensor system including a reader 10 in remotecommunication with a sensor 12 is provided. The reader 10 is capable ofexciting the sensor 12 by transmitting a signal, such as a radiofrequency (“RF”) pulse, at or near the resonant frequency of the sensor12. (See FIG. 1.) The sensor 12 may emit a ring signal for a shortperiod of time in response to the excitation pulse from the reader 10.

The sensor 12 may be a passive device, containing no power source of itsown, and capable of emitting a ring signal 16 in response to anexcitation signal 14 at or near the resonant frequency of the sensor 12.The sensor 12 may be configured to sense a specific parameter. Forexample, the sensor 12 may include a fixed inductor 13 and a capacitor15 that varies based on the sensed parameter. The varying capacitance orinductance alters the resonant frequency of the sensor 12. It should beappreciated, however, that the sensor 12 may be any wireless sensorknown in the art capable of remote communication with the reader 10.Further, while the sensor 12 is described as an RF resonant sensor, itwill be appreciated that the sensor 12 may be an acoustically resonantsensor, optically resonant sensor, or other similar sensor known in theart. The reader 10 may employ corresponding signals to activate thesensor 12. Further, the sensor 12 may be an active sensor or a passivesensor.

In an embodiment, sensor 12 comprises at least one inductive element 13and one capacitive element 15. To vary sensor 12's resonant frequency inproportion to the sensed parameter, either inductive element 13, orcapacitive element 15, or both, may be configured to change inductanceor capacitance proportionately with the sensed parameter. In an exampleembodiment shown in FIG. 1, capacitive element 15 is variable andinductive element 13 is fixed. Typical examples of such components aresensors which change their capacitance in response to changes inpressure. Such capacitive pressure sensors are well known in the art.

In one embodiment, the at least one inductive element 13 in sensor 12also functions as an antenna for sensor 12, coupling energy to and fromanother antenna 26 located on the reader 10.

The reader 10 may excite the sensor 12 by transmitting an excitationpulse 14 in the vicinity of the sensor 12. For example, the reader mayemit a RF excitation pulse 14 at or near the resonant frequency of thesensor 12. The sensor 12 may emit a ring signal 16 in response to theexcitation pulse 14. The reader 10 may determine the frequency of thering signal 16 in order to determine the sensed parameter value.

FIG. 2 is a flow diagram illustrating an example of the steps that maybe involved in the process of the reader 10 acquiring a reading from thesensor 12. Each step may consist of multiple indented steps and suchsteps may be indented several levels. However, only basic, top-levelsteps are shown to clarify the sequence of operation of the readerduring reading acquisition. In the initial condition 202, the sensor 12is already configured such that its resonant frequency is proportionalto the sensed parameter. Some examples of sensed parameters that can bemeasured with capacitive or inductive sensors are pressure, temperature,acceleration, angular rate, PH level, glucose level, salinity,viscosity, dielectric constant, humidity, proximity, electrolyte level,and oxygen level. Additionally, other known parameters may also besensed.

The sensor 12 is located remotely from the reader 10. In one embodiment,the sensor 12 is implanted inside a living human or animal body to takephysiological measurements. Possible locations of interest include, butare not limited to: blood vessels, cranium, eyes, bladder, stomach,lungs, heart, muscle surface, bone surface, or any bodily cavity. Thesensor 12 may be implanted for short-term acute, or long-term chronictime periods. The sensor 12 may be standalone, or may be incorporatedwith another device such as a catheter, stent, shunt, filter, pacemaker,pacemaker wire, vascular closure device, and the like.

The sensor 12 is designed to have an operating frequency range 220 (notshown in FIG. 2) that maps to a range of values of the sensed parameter.When it is desired to acquire a reading, the reader 10 may transmit anexcitation pulse 14 in the vicinity of the sensor 12 as in block 204 ofFIG. 2. The pulse 14 may be a brief burst of energy at a predeterminedfixed frequency. The pulse 14 frequency may be selected to be at or nearthe middle of the sensor 12 operating frequency range 220, and thebandwidth of the pulse 14 may be narrow. An advantage of a narrowbandwidth pulse is that it is less likely to interfereelectromagnetically with other devices around it. A further advantage ofa narrow bandwidth pulse is that it allows the system to comply morereadily with government or industry regulations regardingelectromagnetic spectrum allocation, by enabling system designers toselect a pulse frequency within a tight band specified by suchregulations. In one embodiment, the pulse 14 is narrow and centered at13.56 MHz, which is one of the so-called Industrial, Scientific, andMedical (ISM) bands allocated for use in commercial RF devices by theInternational Telecommunications Union (ITU). Yet another advantage of anarrow bandwidth pulse is that it may require less power than anequivalent continuous-transmit solution, thus making reader 10 moreamenable to battery operation, and allowing the use of smallercomponents which generally require less heatsinking than their higherpowered counterparts. Finally, an advantage of transmitting a fixedfrequency pulse 14 in step 204 of FIG. 2 is that the transmit circuitryof reader 10 is simple compared to swept-frequency orcontinuous-transmit solutions.

Because sensor 12 is in close proximity to reader 10, step 206 of FIG. 2now takes place. Sensor 12 is energized by pulse 14 via inductivecoupling between its antenna and that of reader 10. Pulse 14 causescurrent to flow in the antenna of sensor 12, energizing the ‘LC Tank’circuit formed by capacitor 15 and inductor 13. Pulse 14 is generally ofshort duration, and in step 208, reader 10 abruptly terminates pulse 14.Immediately the energy stored in the LC tank circuit of sensor 12 beginsto dissipate, oscillating at sensor 12 resonant frequency as it does so.The sensor 12 antenna thus emits a ring signal 16 at this frequency.After terminating transmission, the reader 10 must immediately go into areceiving mode, as in step 210, in order to detect ring signal 16 andamplify it.

Depending on measurement conditions, the ring signal may be weak, noisy,or of short duration, leading to accuracy and resolution penaltiesduring frequency measurement. For this reason, the reader 10 may lockand hold the sampled ring signal at constant frequency and strongamplitude in step 212, for a sufficient time to acquire a high accuracyfrequency measurement in step 214.

FIG. 3 illustrates qualitatively the idealized characteristics of thereader 10 and sensor 12 in the frequency domain, in an embodiment.Sensor 12 senses its physical parameter of interest across apredetermined operational range of values. It maps this physicalparameter range onto a corresponding operating frequency range 220.Curve 224 is the transfer function of the sensor 12, when the resonantfrequency of sensor 12 is at the minimum of its operating frequencyrange 220. Sensor transfer function 224 has its peak at the resonantfrequency of sensor 12. As the sensed parameter varies within theoperational range of values, the sensor transfer function movescorrespondingly within operating frequency range 220. Thus, depending onthe value of the sensed physical parameter, the sensor transfer functioncan be centered anywhere within operating frequency range 220. Itsresonant frequency (peak of the transfer function curve) will correspondto the value of the sensed parameter. When the sensed parameter is atthe other extreme of its operational range, the sensor transfer functionbecomes the maximum frequency sensor transfer function 222.

Narrowband function 14 in FIG. 3 represents the excitation pulse 14shown in FIG. 1. Its frequency, designated f_(xmt), is generally fixedto be at or near the center of operating frequency range 220. Pulse 14is generally of narrow bandwidth, short time duration, and is fixed at apredetermined frequency f_(xmt). These pulse characteristics endowreader 10 with several advantages over readers which must sweep or varytheir transmitted frequencies: simpler circuitry, simpler controlsoftware/firmware, lower overall power consumption (enabling batteryoperation), lower power (thus smaller) components, less internal heatdissipation, reduced susceptibility to electromagnetic interference froman outside source, reduced likelihood of interfering electromagneticallywith an outside device, and increased ease of compliance with governmentfrequency allocation regulations.

Another important feature shown in FIG. 3 is the horizontal linerepresenting the minimum signal detection threshold 226 of the reader10. After excitation pulse 14 is switched off, sensor 12 will dissipatethe energy it received from excitation pulse 14. In the absence of theforced excitation pulse 14, this energy causes oscillation at the sensor12 resonant frequency, emitting a ring signal 16 (not shown in FIG. 3).The signal strength (amplitude) of ring signal 16 is determined by theintersection of the excitation pulse 14 and the sensor transferfunction: the ring signal's amplitude will be limited by the product ofthe two functions at that point. The amplitude of this product, at thepoint of intersection, must be greater than or equal to the signaldetection threshold 226 of reader 10 in order for the ring signal 16 tobe detected and measured by the reader 10.

FIG. 4 provides an illustrative example, in the frequency domain, of atypical signal exchange between reader 10 and sensor 12. The processshown in this figure is the same as that shown in flowchart form in FIG.2. In the initial condition shown in FIG. 4a , the sensed parametervalue is such that sensor 12 transfer function 228 is centered at afrequency within the operating frequency range 220. Note that the sensedparameter (and hence transfer function 228) changes on a much slowertimescale than the electronic signals going between sensor 12 and reader10, and hence transfer function 228 is quasistatic relative to thosesignals. Because the sensed parameter is quasistatic related to theelectronic signals, the reader 10 is able to take multiple samples overa short time interval and average those samples to obtain a moreaccurate measurement.

In FIG. 4b , excitation pulse 14 is generated by reader 10. Pulse 14 isa narrow bandwidth signal, centered at frequency f_(xmt), which is at ornear the center of operating frequency range 220. When reader 10generates excitation pulse 14 in the physical vicinity of sensor 12,energy is transferred from reader 10 to sensor 12. In one embodiment,this energy transfer occurs by inductive coupling, with f_(xmt) in theRF frequency band. Note the point of intersection 230 between readerexcitation pulse 14 and sensor transfer function 228. The product of thetwo amplitudes at this point will determine the amplitude of the ringsignal 16.

Next, in FIG. 4c , reader 10 stops transmitting excitation pulse 14.When excitation energy ceases, the sensor 12 shifts from a forced drivecharacteristic at the transmit frequency with phase error due tooff-transmit-frequency resonance, to a passive resonant characteristicat a frequency dependent on the resonant frequency of the sensor and itssurroundings, approximately at the peak of curve 228. Due to resonantenergy within the inductor of the sensor 12, a time-varying magneticfield is generated around the sensor 12 at this resonant frequency,which can be detected at the reader 10 as an emitted signal at thisresonant frequency.

Note that if sensor 12 is exposed to a sensed parameter that movestransfer function 228 still further to the right in FIG. 4b (in thedirection of increasing f_(res)) then the amplitude of the curve 228 atthe point of f_(xmt) decreases, causing intersection level 230 todecrease as well. As f_(res) increases further and reaches f_(max),intersection amplitude 230 equals reader 10's minimum detectionthreshold 226. If transfer function 228 moves still further to theright, f_(res) exceeds f_(max), and the intersection amplitude 230 fallsbelow the detection threshold 226 of reader 10. Now reader 10 can nolonger detect the ring signal 16, i.e. f_(res) is outside the operatingfrequency range 220 of the system. Note that sensor 12 must be designedsuch that its transfer function 228 has wide enough bandwidth tomaintain an intersection amplitude 230 above the detection threshold 226of reader 10 across the entire operating frequency range 220. However,designing sensor 12 with a wide transfer function 228 generally lowersthe peak amplitude of transfer function 228, so a balance must be foundbetween amplitude and bandwidth. In general, it is clear from FIG. 4that the reader 10's ability to detect and measure ring signal 16 willalso depend on the power level of the ring signal after cessation ofexcitation pulse 14, on the system Q, and the time duration of ringsignal 16.

The shapes of transfer function 228, signals 14 and 16, and theoperating range 220 shown in FIG. 4 are illustrated as examples. In someembodiments, transfer function 228 may have different characteristics,and may not be symmetrical about f_(res), which is at its peak.Additionally, operating range 220 may not be symmetrical about f_(xmt),the frequency of excitation pulse 16. Operating range 220 asymmetry mayoccur as a result of the sensor 12 characteristics, or may be purposelydesigned in, in order to offset asymmetries in transfer function 228,excitation signal 16, or ring signal 14.

In an alternate embodiment, reader 10 may transmit a pulse which is notnear the center of the sensor 12 operating range 220. In this casereader 10 transmits a pulse at a frequency that is harmonically relatedto a frequency inside operating range 220 of sensor 12. That is, ahigher or lower harmonic resulting from the transmitted pulse or pulsesis used as the excitation pulse 16 shown in FIG. 4.

In yet another embodiment, reader 10 may transmit two or more excitationpulses at different frequencies, either simultaneously or at differenttimes. These multiple excitation pulses may excite different parts ofthe operating frequency range 220. Alternatively, frequencies created byadding or subtracting combinations of these multiple pulses, or theirharmonics, may serve as the excitation frequency 16 in FIG. 4.Excitation pulses may also assume a Gaussian, or other non-sinusoidalshape.

Referring again to FIG. 1, reader 10 may also incorporate circuitry forconverting the ring frequency readings from sensor 12 to digital form,and storing these in an on-board memory. Besides measurements fromsensor 12, reader 10's memory may also store other relevant data.Examples include timestamp data, calibration coefficients, firmwarerequired to accomplish system functions, firmware upgrades, partnumbers, serial numbers, usage logs, historical data, configurationdata, diagnostic data, information about the host location andapplication of the sensor, and user-defined data.

Reader 10 may also incorporate human interfaces such as a displayscreen, LEDs, or an audible indication, corresponding to some aspect ofthe frequency data. Further, reader 10 may process the frequency data itreceives, performing such functions as averaging, filtering,curve-fitting, threshold monitoring, timestamping, trend analysis,comparison with other data, and the like.

Reader 10 may also communicate with a data interface 17, as shown inFIG. 5. Data interface 17 is external to reader 10, and is configured toreceive electronic signals from reader 10, and transmit signals toreader 10. Additionally, data interface 17 may provide power to reader10, for example charging a battery located in reader 10. Examples ofdata interface 17 include a host computer, a docking station, atelephone network, a cell phone network, a GPS network, an opticalnetwork, a Bluetooth network, a storage area network, an internetwebsite, a remote database, a data input device, an audible sound, and adisplay screen.

The reader 10 and data interface 17 may be connected directly to oneanother or indirectly through an intermediate device, or may communicatevia a remote connection. They may reside in the same housing. The reader10 and data interface 17 may be connected via a cable or by a wirelesslink. The reader 10 may send information to the data interface 17.Examples include data related to the sensor 12, measurements taken fromsensor 12, timestamp data, part number, serial number, firmware revisioninformation, usage logs, diagnostic data, historical data, status data,configuration data, information about the host location and applicationof the sensor, and user-defined data. The data interface 17 may providedata and commands to the reader 10. For example, the data interface 17may provide reader 10 with information regarding schedules and intervalsfor sampling the sensor 12, calibration coefficients or lookup tables,firmware required to accomplish system functions, firmware upgrades,configuration settings, diagnostic commands, resets, restarts,user-defined data, and user-issued commands.

The data interface 17 may further communicate with a remote data system18 to exchange status and control signals, as well as provide sensordata. The remote data system 18 may include a data gathering module 19to receive data from the data interface 17, a data logging module 20 tostore the received data, and a data display 21 to display the sensordata. Like the data interface 17, the remote data system 18 may storeand process the data, issue commands, and distribute these data andcommands, allowing communication with multiple users over a datanetwork. Like the connection between reader 10 and data interface 17,the connection between data interface 17 and remote data system 18 maybe through a cable or may be wireless. The configuration shown in FIG.5, where the reader 10 connects to data interface 17 through a cable,and data interface 17 connects to remote data system 18 wirelessly, isone example embodiment. Although the example in FIG. 5 associates thefunctions of data logging and display with remote data system 18, itwill be obvious to those of normal skill in the art that these functionsmay also be carried out by external data interface 17 or reader 10.

The system of reader 10, sensor 12, and data interface 17 describedabove is particularly advantageous in one embodiment in the field ofbiomedical telemetry. In this embodiment sensor 12 is implanted into aliving human being, to sense a physiological parameter, for exampleblood pressure sensed from within an artery. Sensor 12 is well-suitedfor this application as it can be made very small by conventionaltechniques, and as it is a passive sensor it requires no on-board powersource that will eventually be exhausted. Reader 10, for its part, canbe physically small enough to be handheld, battery-powered, thermallycool, and electromagnetically compatible with other electronics in itsvicinity. These attributes stem from the simple, low-power circuits thatgenerate the narrowband, fixed frequency excitation pulse 14 asdescribed above. Thus reader 10 may be worn comfortably on a person'sclothing in the vicinity of the implanted sensor 12, taking frequentreadings and processing/storing them. Periodically, for example daily,the user may place reader 12 on data interface 17 in the form of adocking station. Data interface 17 may contain circuitry to charge thereader 12 battery, update reader 12 settings and software, and downloadits data. Data interface 17 may also communicate this data to the user,and other interested persons such as the user's physician, via aninternet or telephone link. Because of the low-power excitation schemeused by reader 12, such a system can take frequent, accurate bloodpressure readings with a minimum of inconvenience to a patient, andcommunicate these to caregivers efficiently. Clearly, this embodiment isalso applicable to sensing any other internal physiological parameterwhich can effect a change in resonant frequency on a passive LC sensor.

In a variation of this embodiment, sensor 12 is incorporated withanother implantable medical device that performs a different function.For example, sensor 12 may be a blood pressure sensor incorporated witha vascular closure device, such as the Angio Seal product from St. JudeMedical, Inc, of St. Paul, Minn. In yet another variation of thisembodiment, reader 10 may be incorporated with another device. Forexample, reader 10 may be attached to a cell phone, a pair of glasses, ahandheld music player, a video game, an article of clothing, or awristwatch.

Sensor 12, comprising capacitor 15 and inductor 13, may be such thatthese circuit elements are assembled in a single package. Alternatively,some applications may make it advantageous to locate capacitor 15 awayfrom inductor 13, with the two elements connected by conductive leads.As an example, in the embodiment where sensor 12 is implanted in a humanbody, the pressure-sensitive capacitor 15 might be located at the sitewhere the pressure of interest is found, and the inductor 13, which actsas an antenna, might be located closer to the skin surface, minimizingthe wireless coupling distance between sensor 12 and reader 10. Theconnecting conductive leads may take any of a number of well-knownforms, including wires, wire filaments, printed flex circuits, printedrigid circuits, feedthroughs, or rigid pins.

In the implantable embodiment, it may also be advantageous to designsensor 12 to be amenable to minimally invasive implant methods, such ascatheter-based delivery. Additionally, it may be desirable for a portionof the implantable sensor to be radio-opaque or ultrasound-reflective,to aid implant and post-implant diagnostics.

Sensor 12 can be manufactured by a number of well-known technologies.Capacitive sensor 15 may be manufactured by microelectromechanicalsystems (MEMS) technology, lithographic techniques, or classic machiningtechniques. Inductor 13 may be a wirewound coil; an FR4, Teflon, Rogers,or other printed circuit board; a Low Temperature Cofired Ceramic(LTCC), greentape, or other ceramic printed circuit board;. or any otherinductor technology known to those in the art. Inductor 13 may be coredor non-cored, and may further utilize magnetic materials incorporatedinto one of the printed circuit board or ceramic technologies mentionedabove. The inductor and capacitor may be packaged together as amulti-chip module (MCM).

In another embodiment, the system of FIG. 1 may further comprise anintermediate antenna 240, as shown in FIG. 6. Intermediate antenna 240comprises two antennas: reader-side antenna 242 and sensor-side antenna244, which are connected together in series. The intermediate antenna240 may improve signal coupling between reader 10 and sensor 12, and maybe useful in cases where there are multiple barriers 246 and 248 betweenthe reader 10 and sensor 12, which are not easily penetrated byconductive leads. As an example, for a sensor 12 implanted in a bloodvessel, Barrier 2 (248) represents the blood vessel wall, and Barrier 1(246) represents the skin surface. With the intermediate antenna 240 inplace, signal coupling between reader 10 and sensor 12 is moreefficient, as it takes place by conduction through leads rather than byradiation through whatever medium the system is in. Additionally, theantennas 242 and 244 can each be sized to match their correspondingantennas on sensor 12 and reader 10, further improving couplingefficiency. Finally, the sensor side antenna 244 can be aligned withprecision across from sensor inductor 13, reducing errors due tomisalignment between reader 10 and sensor 12 that might occur in theabsence of the intermediate antenna 240. The intermediate antenna 240can be made from flex circuits, wirewound coils, or other widelyavailable means. Note too that the concept can be extended toapplications where more than two barriers exist, by adding moreintermediate antennas 240 for each pair of barriers.

In another embodiment, the sensor 12 in FIG. 1 may further comprise asecond LC Tank circuit, with a separate inductor and capacitor, calledthe Reference Resonator. The Reference Resonator may be fabricated usingthe same materials, processes, and parts as the Sensing Resonatorcomprised of inductor 13 and capacitor 15, but with two key differences.First, the Reference Resonator's components are fixed in value and donot vary with the sensed parameter. Second, their fixed resonantfrequency is designed to be outside the Operational Frequency Range 220of the sensing resonator. The purpose of the Reference Resonator is toprovide a background reading which can be used to correct the sensorreading acquired by the reader 12. Certain factors that lead toinaccuracy, such as reader distance, changes in the intervening medium,sensor orientation to the reader, aging of components, mechanicalstress, electrical stress, outgassing, temperature, cell growth, bloodclotting, etc, may affect the Reference Resonator in a manner similar tothe sensing resonator. By understanding the relationship betweenReference Resonator deviation from its fixed frequency and sensingresonator deviation from its nominal frequency, the reader can providecorrection factors to the sensed frequency based on the Referencereading. In this embodiment, extra steps are introduced into FIG. 2between steps 202 and 204, in which the reader 10 transmits anexcitation pulse at the Reference Resonator's nominal resonantfrequency, observes any deviation in Reference ring frequency, andcalculates (or obtains from a lookup table) an appropriate correctionfactor for the forthcoming reading obtained in step 210. Alternatively,the reference reading can be taken after the sense reading. Althoughevery change the Sensing Resonator undergoes may not affect theReference Resonator in exactly the same way, this method of“self-calibration” may improve performance by eliminating or reducingsome of the inaccuracies that are common to both resonators. These maybe, for example, associated with distance, orientation, physiologicalreactions, changes in intervening tissue, and other long-term changes insensor 12 behavior often referred to collectively as “sensor drift”.Additionally, care must be taken in the frequency selection, and otherdesign aspects of the Reference Resonator, to avoid coupling with theoriginal sensing resonator, and common interaction with the reader.

The reader 10 includes circuitry to send the excitation pulse 14,receive the ring signal 16, and process the ring signal 16. (FIG. 7.)For example, the reader 10 includes timing and control circuitry 22 toconfigure and activate the other circuits in the reader 10. The solidarrows to and from the timing and control circuitry 22 represent thecontrol interfaces, such as digital or low-frequency signals. The timingand control circuitry 22 further generates an RF signal (illustrated asthe broken line arrow) that is sent to transmit circuitry 24. Thetransmit circuitry 24 receives the RF signal and sends the excitationpulse 14 to antenna 26 to excite the sensor 12. The timing and controlcircuitry 22 may only provide the RF signal to the transmit circuitry 24during the intervals when the excitation pulse is being transmitted toprevent leakage or coupling to other nodes in the system.

The reader 10 further includes an antenna 26 connected to the transmitcircuitry 24 and the receive circuitry 28. The transmit circuitry 24utilizes the antenna 26 for transmitting the excitation pulse 14, whilethe receive circuitry 28 utilizes the antenna 26 for receiving the ringsignal 16. In an embodiment, the antenna 26 is connected to both thetransmit circuitry 24 and the receive circuitry 28 at all times insteadof being switched between transmit and receive. This shared antenna 26design requires special consideration to prevent damage to the receivecircuitry 28. Specifically, care must be taken not to overload thesensitive amplifier stages of the receive circuitry 28. Additionally,the reader 10 requires a fast transition between the extreme overdrivecondition present while the transmit circuitry 24 is driving the antenna26, and the low voltage condition present at the antenna 26 during thereceive and amplify phases. For instance, the voltage at the antenna 26may exceed 200 volts peak-to-peak during transmission of the excitationpulse, and may be single-digit millivolts, decaying rapidly tomicro-volts, during reception immediately following the excitation pulse14. While the reader 10 is described as having a shared antenna 26, itwill be appreciated that the reader 10 may incorporate more than oneantenna to separately perform the functions of transmitting theexcitation pulse 14 and receiving the ring signal 16.

The reader 10 further includes a phase locked loop (PLL) 30 to receiveand lock onto the ring signal 16. The receive circuitry 28 may amplifyand condition the ring signal 16 before sending it to the PLL 30. ThePLL 30 includes a voltage controlled oscillator (“VCO”) 32 (not shown inFIG. 7) that may operate to lock a frequency within the range of sensorresonance frequencies when no signal is present, or may be chosen toprefer a frequency above or below the range of sensor resonancefrequencies when no signal is present to enhance lock time when a sensorresonance frequency is received. In an embodiment, a PLL was chosen thatperformed better when the no-signal PLL lock frequency was slightlyabove the range of sensor resonant frequencies. VCO 32 generates an acsignal which is proportional to the ring signal frequency, called thecount signal 250. The PLL 30 adjusts the divided-down count signal tomatch the ring signal 16 frequency, and sends the count signal 250 to acounter 34. The VCO 32 interfaces with frequency counter 34 whichdetermines the count signal 250 frequency, and provides a digital signalrepresenting that frequency to external interface circuitry 36 fortransfer to the data interface 17. By operating the VCO 32 at a higherfrequency than the ring signal 16, the time required to count and recordthe VCO 32 count signal 250 frequency may be significantly decreased.

Each component of the reader 10 is designed to operate efficiently andreduce power consumption. To that end, the reader 10 includes a reducedpower functionality. The timing and control circuitry 22 controls thepower status of each component by way of a wakeup timer 38 connected toeach component. (FIG. 8.) In reduced power mode, some components may becompletely powered down while other components may operate in a sleepmode, where power remains to maintain configuration but the circuitbecomes static to minimize power consumption.

The timing and control circuitry 22 may place each component of thereader 10 in a sleep or powered-down mode when not in use. Additionally,the entire reader 10 may be placed in a low-power mode at the systemlevel for periods of time specified by an external controller. Thetiming and control circuitry 22 may include a configuration buffer 40that receives timing instructions from the external interface circuitry36. The instructions establish the timing period before entering intoreduced power mode, and other timing periods for the wakeup timer 38.Besides timing instructions from outside the reader 10, entry into/exitfrom reduced power mode may also be triggered by a threshold beingexceeded for one of the on-board signals. The firmware of reader 10 maycontain algorithms for deciding to enter/exit reduced power mode.

During a reading acquisition, the wakeup timer 38 may wake up eachcomponent of the reader 10 at the appropriate time to ensure that eachcomponent is in an operational state when needed. Specifically, thewakeup timer 38 may communicate with a transmit timer 42, a receivetimer 46, a PLL timer 48, and a frequency counter timer 50 to wake upand control the respective components of the reader 10. Once initiated,each of these timers may control and power up the respective component.When configured, the wakeup timer 38 may delay for a specified interval,which may be zero seconds, before sending an initiate signal 52 to startthe other timers. As illustrated in FIG. 8, the initiate signal 52 isnot shown as a continuous line from the wakeup timer 38 to therespective timers in order to prevent line crossings and minimizeconfusion.

Once initiated, the transmit timer 42 establishes proper sequence andperiod to the power control 54, damp control 56, Q control 58, and RFenable 60 signals to properly sequence the transmit circuitry 24 andtransmit frequency generator 44. The power control signal 54 controlsthe power status and sleep status of the transmit circuitry 24. The dampcontrol signal 56 controls the activation of a damping circuit in thetransmit circuitry 24 to quickly dissipate antenna 26 energy at the endof a transmission period. The Q control signal 58 controls a switchingcircuit in the transmit circuitry 24 to reduce the Q and modify thebandwidth of the antenna 26 during reception of the ring signal 16. TheRF enable signal allows the transmit frequency generator 44 to send anRF signal to the transmit circuitry 24. In an embodiment, the transmitfrequency generator 44 only provides the RF signal to the transmitcircuitry 24 during periods where the transmit circuitry 24 istransmitting an excitation pulse 14.

The receive timer 46 is configured to establish proper sequence andperiod to the power control signal 62 to properly sequence the receivecircuitry 28.

The PLL timer 48 establishes proper sequence and period to the powercontrol 64 and S/H mode 66 signals to properly sequence the PLL 30. Thepower control signal 64 controls the power status and sleep status ofthe PLL 30. The S/H mode signal 66 controls a sample and hold circuit inthe PLL 30, used to cause the PLL to lock onto the transmitted frequencythen onto the ring signal 16 frequency, then hold the VCO 32 countsignal 250 frequency at the locked frequency until that frequency ismeasured by the counter 34.

The frequency counter timer 50 establishes proper sequence and countinterval to the power control 68 and start/stop count 70 signals toproperly sequence the frequency counter 34. The power control signal 68controls the power status and sleep status of the frequency counter 34.The start/stop count signal 70 controls the time that the frequencycounter 34 begins and ends measuring the VCO 32 count signal 250frequency.

Note that although FIG. 8 contains signals that share the same name,such as ‘Initiate’, ‘Configure’, and ‘Power Control’, each of thesesignals is unique to the circuit block it connects to. For example,power control signal 68 from Frequency Counter Timer block 50 is not thesame signal as power control signal 64 from PLL Timer block 48, asdescribed above.

The transmit circuitry 24 is configured to transmit the excitation pulse14 to the sensor 12 by way of the antenna 26. (FIG. 7.) The excitationpulse 14 may be a fixed or rapidly varying frequency burst at or nearthe resonant frequency of the sensor 12. For example, the excitationpulse 14 may be a fixed frequency burst within several bandwidths of thesensor 12 resonant frequency. Alternatively, the excitation pulse 14 maybe a fixed or rapidly varying frequency burst or sweep of a very shortduration, at or near a frequency harmonically related to the sensor 12resonant frequency. The excitation pulse 14 may also be an ultra-wideband pulse. This plurality of excitation pulse 14 approaches is possiblebecause the ring signal 16 is received when the excitation pulse 14transmissions have ceased. Therefore, excitation pulse 14 transmissionsmay be limited to frequency bands, amplitudes, and modulation schemesacceptable to regulatory government bodies. Radio frequency regulationsmay not apply to the sensor 12 as the sensor 12 is a purely passivedevice.

The excitation pulse 14 does not require significant transmission timebecause a single short transmission of energy results in a single andcomplete sample of the ring signal 16. Power consumption may be reducedby using a lower transmission duty cycle, thereby reducing the dutycycle of transmit, receive, counting, and digital processing circuitry.By reducing power consumption battery power becomes a much more viableoption to power the reader 10.

The excitation pulse 14 may be configured to maximize several systemparameters. For example, if a fixed frequency excitation pulse 14 isused, the frequency of the burst may be configured to maximizeparameters such as maximum allowable transmit peak power, maximumfreedom from in-band or near-band interference during the “receive”interval while the PLL is being locked to the ring signal 16, maximumworldwide acceptance of a particular frequency for reader transmissionsfor the desired sensor purpose, or other such criteria.

FIG. 9 illustrates the transmit circuitry 24. A level shifter 72 of thetransmit circuitry 24 receives control signals 54, 56, 58 and the RFsignal from the timing and control circuitry 22. The level shifter 72buffers the inputs and converts control logic levels to circuit drivelevels. A transmit driver 74 amplifies the RF signal to providesufficient power to drive the antenna 26. The Q control circuit 76 isactivated during receive to reduce the Q of the combined antenna 26 andtuning and D.C. block 82. A damping circuit 78 is briefly activatedimmediately at the end of transmission of the excitation pulse 14 toabsorb energy in the antenna and allow the antenna to respond to thering signal 16. The damping circuit 78 may provide a different Q factorto the antenna to improve reception of the ring signal 16. The powercontrol circuitry 80 controls the power-on and sleep mode for componentsin the transmit circuitry 24. The tuning and D.C. block 82 adjuststuning for the antenna 26 and prevents direct current from improperlybiasing the damping circuit 78. The RF output or excitation pulse 14from the transmit circuitry is routed to both the antenna 26 and thereceive circuitry 28.

Once the excitation pulse 14 is transmitted by the transmit circuitry24, the receive circuitry 28 is configured to listen for the ring signal16. With reference to FIG. 10, a high Z buffer/clamp 84 includes a highimpedance (“high Z”) input device that limits the effect of the receivecircuitry 28 on the tuning performed by the tuning and D.C. block 82.The high Z buffer/clamp 84 further serves to protect the amplifierstages 86 from the extreme voltages present on the antenna 26 duringtransmission of the excitation pulse 14. Voltages at the antenna 26 mayreach upwards of 200 volts peak-to-peak during transmission of theexcitation pulse, requiring only approximately 60 pico-farads ofcapacitance to tune the antenna 26. In an embodiment, a 1 pico-faradcapacitor is used as a high impedance input current limiting device on a13.56 MHz transmit circuit. Low capacitance diode junctions that shuntover-voltage to the power supply and undervoltage to ground may beplaced on the receiver side of the 1 pF capacitor, so that the capacitorlimits current through the diodes as they protect the receiver amplifierfrom high voltages during transmission through the antenna 26.

The amplifier stages 86 amplify the ring signal 16 to a sufficient levelto drive the PLL 30 input. Careful design of the amplifier stages 86 isrequired to achieve adequate transient response when the transmittedexcitation pulse 14 signal is removed and damped, and the low level ringsignal 16 is received. Common gate amplifier stages with low Q tunedreactive drain loads may be used to condition the high Z buffer/clamp 84output, followed by several filters interspersed between high gainamplifier stages. The filters may be either resistor-capacitor (“RC”)filters or inductor-capacitor (“LC”) filters. In an embodiment, thefilters may all be RC bandpass filters. Another common gate amplifierstage with low Q tuned reactive drain load may be used for finalbandpass conditioning prior to feeding the signal to the PLL 30 input.This design enables all of these amplifier types to perform fromextremely low signal input levels to extremely high signal input levelswithout signal distortion such as frequency doubling or halving due tostage saturation characteristics, as well as the excellent high inputimpedance achievable with the common-gate amplifier stages and theoutstanding transient response characteristics of the RC filterinterspersed between high gain amplifier stages. Special care must betaken in stage-to-stage power and signal isolation to prevent unwantedoscillations due to the extreme gain associated with the amplifierstages 86.

Power control circuitry 88 may apply and remove power to and from theamplifier stages 86 and the buffer in the high Z buffer/clamp 84 toreduce power consumption. It should be noted that the high Zbuffer/clamp 84 is designed to provide full protection even with powerremoved as excess energy will merely power up the amplifier stages 86until dissipated. The input impedance is high enough to limit excessenergy to prevent overpowering the amplifier stages 86. In anembodiment, the receive circuitry 28 is active during the transmissionof the excitation pulse 14 to decrease the time required for the PLL 30to lock onto the ring signal 16.

The PLL 30 receives the amplified and conditioned ring signal 16 fromthe receive circuitry 28. With reference to FIGS. 10 and 11, the RFsignal from the receive circuitry 28 amplifier stages 86 feeds an RFbuffer 90 of the PLL 30. The RF buffer 90 may feed the RF signal to anoptional RF divider 92 that divides the RF signal frequency by aninteger value. (FIG. 11.) The RF divider 92 then feeds the RF signal toa first input of a phase frequency detector 94. The output of thefrequency detector 94 feeds a sample-and-hold (S/H) error amplifier 96.The S/H error amplifier 96 controls the frequency of the VCO 32. Thecount signal 250 output by the VCO 32 feeds the VCO divider 98, whichoutput in turn feeds a second input to the phase frequency detector 94.The PLL 30 may include an output buffer 102 to reduce loading of the VCO32 while forwarding the count signal 250 frequency to the frequencycounter 34. The VCO divider 98 allows the VCO 32 to operate at afrequency significantly higher than the ring frequency 16. As a result,the time required to count and record the VCO signal frequency may besignificantly reduced. Moreover, the shorter count interval reduces VCOdrift during counting and allows a higher sample rate.

The phase frequency detector 94 is configured to determine the frequencyand phase error between the divided RF signal and the divided VCOsignal. This is best accomplished by filtering and amplifying the signalthat is fed to the S/H error amplifier 96. Further, the S/H feature mayoptimally forward the filtered and amplified signal to control the VCO32. In this manner, a closed control loop is formed that causes the VCO32 count signal 250 frequency to equal the ring signal 16 frequencytimes the VCO divider 98 integer divided by the RF divider 92 integer.The PLL 30 may include additional frequency dividers to optimize thecircuit design and increase the potential VCO 32 frequency range.

The PLL timer 48 sends a S/H mode control signal 66 to the S/H erroramplifier 96 of the PLL 30. The S/H mode control signal 66 may place theVCO 32 in a sample mode. In an embodiment, the VCO 32 is placed insample mode for a predetermined length of time. In sample mode, thedivided VCO count signal frequency is adjusted to match the ring signal16 frequency, as described above. When the S/H mode control signal 66 isplaced in the hold mode, the S/H error amplifier 96 will hold its outputconstant, causing the control voltage to the VCO 32 to be approximatelyconstant over a length of time sufficient to determine the VCO 32 countsignal 250 frequency.

The power control signal 64 from the PLL timer 48 to the power controlcircuitry 104 determines whether the PLL 30 is in a power on or asleep/power-off mode to conserve electrical power. Depending on thespecific PLL 30 that is used, a control and communication link (notshown) may be required to set the RF divider 92 integer, the VCO divider98 integer, and the phase frequency detector 94 outputs and outputconfigurations. The communications link may be specific to theparticular PLL 30 used.

The frequency counter 34 includes counter stages 106, a counter buffer108, and a power control circuitry 110 as shown in FIG. 12. Thefrequency counter timer 50 sends a start/stop control input 70 to thecounter stages 106 and counter buffer 108. The frequency counter timer50 also sends a power control input 68 to the power control circuitry110. The counter stages 106 count the VCO signal frequency from the PLL30 output buffer 102. The counter stages 106 start counting when thestart/stop control commands start, and end when the start/stop controlcommands stop. When the start/stop control commands stop, the counterbuffer 108 is loaded with the count value from the counter stages 106.The power control circuitry 110 controls the power-on and sleep modesfor components in the frequency counter 34. The counter buffer 108output may supply a count input to the external interface circuitry 36.The ring frequency 16, and subsequently the sensed parameter, may bedetermined from the frequency count.

In other embodiments, other methods for measuring the received andamplified frequency are possible. These may include direct counting ofthe ring signal, or various frequency-to-voltage conversion circuitsknown in the art.

In operation, the reader 10 sequences as follows. During periods of timewhen the sensor 12 is not being sampled, all components of the reader 10are placed in reduced power mode. The wakeup timer 38 in the timing andcontrol circuitry 22 is configured for a particular sample delay orsample interval. At the specified time, the wakeup timer 38 initiates asample sequence. Specifically, the wakeup timer 38 powers up or wakes upeach component of the reader at appropriate times to ensure eachcomponent is in an operational state when needed.

The external interface circuitry 36 is generally not required in thesampling sequence, except to receive the final data generated. Its entryinto/exit from low power mode may be handled by internal or externalcontrollers other than timing and control circuitry 22. The timing andcontrol circuitry 22 provides the RF signal to the transmit circuitry 24for a short period of time, such as approximately 20 microseconds. TheRF signal from the timing and control circuitry 22 is then terminatedand the transmit circuitry 24 is controlled to dampen the transmittedsignal at the antenna 26 quickly. The transmit circuitry 24 is thenplaced in an appropriate mode to allow reception of the ring signal 16at the antenna 26. In an embodiment, when the antenna 26 is configuredto receive the ring signal 16, the antenna 26 damping is greater thanthe ring signal 16 damping.

During transmission of the excitation pulse 14, the receive circuitry 28receives, conditions, and clamps the transmitted RF signal at theantenna 26. Once transmission of the excitation pulse 14 ceases and theantenna 26 is configured to receive the ring signal 16, the receivecircuitry 28 transitions into a high-gain reception mode to receive thering signal 16 from the antenna 26. The PLL 30 is in sample mode toallow the RF buffer 90 to receive the conditioned output of the receivecircuitry 28. When the antenna 26 begins to receive the ring signal 16,the PLL 30 shifts from locking onto the transmitted excitation pulse 14frequency, to locking onto the ring signal 16 frequency. After a timeinterval sufficient for the PLL 30 to lock onto the ring signal 16frequency, the PLL 30 is shifted to hold mode to maintain VCO 32 countsignal 250 frequency at ring signal 16 frequency. The time required tolock may be predetermined, or may be adaptive base on detected PLLlocked conditions. After lock, the receive circuitry 28 and transmitcircuitry 24 are powered down or placed in sleep mode as appropriate.

Once the PLL 30 is in hold mode, the timing and control circuitry 22instructs the frequency counter 34 to conduct a controlled intervalcount of the VCO 32 count signal 250 frequency. Upon completion of thecount, the PLL 30 components are powered down or placed in sleep mode asappropriate and the count value is transferred to the external interfacecircuitry 36. The frequency counter 34 components are then powered downor placed in sleep mode as appropriate, and subsequently the timing andcontrol circuitry 22 components are powered down or placed in sleep modeas appropriate. If programmed for interval sampling, the timing andcontrol circuitry 22 wakeup timer 38 counts until the next sample isdue. Otherwise, the timing and control circuitry 22 awaits a wakeupcommand with any other needed instructions from the external interfacecircuitry 36. In burst sampling modes, the power up time needed forcomponents to be ready may precede the power down time, in which casethe components would remain powered up until completion of the sampleburst.

An embodiment of the PLL circuit 30 in reader 10, shown in FIG. 13,includes several features that may be added to the PLL 30 to achievealternate but equivalent functionality from the PLL 30 circuit describedabove. Some or all of the changes seen between FIG. 11 and FIG. 13 maybe applied to enhance the operation of the FIG. 11 PLL 30. Theselectable input RF Buffer 111 allows either the RF signal fromamplifier stages 86, or a reference signal generated elsewhere in thereader 10, to be selected for input to RF Divider 92. The selection isdetermined by RF Buffer 111's reference/receive control input. The erroramplifier 112 has been simplified and no longer provides directly thesample and hold capability previously described for the S/H erroramplifier 96 from FIG. 11.

Circuit elements including an analog to digital (A/D) converter 113, adigital to analog (D/A) converter 114, and a switch 115, are illustratedin FIG. 13. These elements may be used to achieve the sample and holdfeature. In the FIG. 13 configuration, a reference frequency signal “RefSignal” may be selected as input to RF Buffer 111 during the reader 10excitation pulse 14 transmission to the sensor 12, and the referencesignal maintained until such time as the RF signal at In A of theselectable input RF buffer 111 becomes stable and available from thereceive circuitry 28. This reference signal allows the PLL 30 to“pre-lock” on a stable reference signal, so reducing lock time when aring signal becomes available from the receive circuitry 28. The outputof the selectable input RF buffer is divided by any value equal to orlarger than 1 by the RF divider 92, then the divided buffer signal isfed into the phase frequency detector 94. The output of the phasefrequency detector 94 feeds an error amplifier 112 that provides theproper gain and frequency response needed to act as the control signalto the VCO 32 in the PLL 30. The output of the error amplifier 112 feedsinput A of the switch 115. When selected to input A, the switch 115passes the error amplifier 112 signal to both the VCO 32 and the A/Dconverter 113. The A/D converter 113 is then used to sample the controlvoltage to the VCO to determine the control voltage level at which theVCO 32 is locked at a frequency related to input A of the selectableinput RF buffer 111. The A/D converter 113 signal may be used to measurethe VCO 32 frequency indirectly as will be described later, and may beused to determine an appropriate setting for the D/A converter 114 suchthat the switch 115 can be set to input B to maintain the VCO 32 at thelocked frequency input level for any period of time, achieving a digitalsample and hold feature similar to that described for the S/H erroramplifier 96 in FIG. 11.

Several slight modifications to the described operation of the FIG. 13circuit may allow functionally equivalent results. One such modificationis calibration of the A/D converter 113 voltage to specific receivecircuitry 28 RF signal frequencies using input B of the selectable inputRF buffer 111 fed with known frequencies. Once calibrated so that therelationship between signal input to RF buffer and the digital output ofthe A/D Converter 113 is well defined, the A/D Converter 113 output canbe used to represent the ring signal 16 frequency. The A/D Converter 113output becomes the PLL output. Operation in this manner will allow theA/D converter 113 to partially or completely supplant the functionalityof the output buffer 102 and frequency counter 34.

Another modification in the described operation of the FIG. 13 circuitis to use the data from the A/D converter 113 for lock analysis of thePLL 30 to reduce lock time and improve lock frequency accuracy. This ispossible because the output of the error amplifier 112 will convergeupon the lock voltage value when the sensor 12 signal 16 is available atthe output of the receive circuitry 28, then will diverge in apredictable manner when the sensor 12 signal 16 level decays past wherelock can be maintained.

Another modification in the described operation of FIG. 13 circuit is touse the D/A converter 114 to generate specific voltages at the VCO 32input, recording the A/D converter output at these specific voltages,and determining the frequency of the signal at the output of the outputbuffer 102, using the frequency counter 34. This allows calibration ofthe A/D converter for one or more frequencies using the frequencycounter 34.

Minor modifications of FIG. 13 circuit that should be obvious to one ofnormal skill in electronic design include rearranging the location ofthe switch 115 and D/A converter 114 from the FIG. 13 shown location tobetween the phase frequency detector 94 and the error amplifier 112.This arrangement requires the additional step of calibration of the D/Aconverter 114 output through the error amplifier 112 to determine properscaling to achieve a desired VCO 32 control voltage, done using eitherthe A/D converter 113 or the frequency counter 34, or both. However,this arrangement allows the D/A converter 114 to be used for pre-lockinstead of using the reference signal at input B of the selectable inputRF buffer 111. This arrangement in combination with the A/D converter113 calibration scheme previously described as allowing elimination ofthe output buffer 102 and the frequency counter 34, may allow formoderate reductions in power required to operate the reader 10 byshortening the time required to resolve the sensor 12 resonant frequencyfor each ring cycle. Another minor modification of the describedembodiment is to distribute the system processing load in appropriatelocations based on power limitations, computational complexity, timecritical requirements, or other system-related priorities. Such amodification might lead a designer to place processing or analysis ofdata from the A/D converter 113, for the D/A converter 114, or thefrequency counter 34 in any of the remote data system 18, the reader 10,or the external data interface 17.

In yet another embodiment of reader 10 circuitry, digital spectrumanalysis circuitry replaces PLL 30 and Frequency Counter 34 in FIG. 7,resulting in the modified block diagram shown in FIG. 14. Here DigitalSampling Circuitry 260 replaces PLL 30, and Spectrum Analysis Circuitry262 replaces Frequency Counter 34. Analog Count signal 250 is likewisereplaced by Digital Count Signal 264.

Functionally, digital sampling circuitry 260 extracts and digitizesinformation from the ring signal 16 during its short ring duration. Thereceive circuitry 28 may amplify and condition the ring signal 16 beforesending it to the digital sampling circuitry 260. The digital samplingcircuitry 260 may directly sample the radio frequency output of thereceive circuitry 28 to obtain time-domain based data for furtheranalysis.

In an embodiment, the reader 10 further contains spectrum analysiscircuitry 262 for converting the time domain data output from thedigital sampling circuitry 260 into frequency domain data, and forbuffering the frequency domain data for forwarding to external interfacecircuitry 36. The spectrum analysis circuitry 262 may also includediscrimination functionality to determine the ring frequency for thering signal 16. As will be obvious to those skilled in the art, some orall of the spectrum analysis circuitry 262 functionality may be readilycarried out by the reader 10 or by the remote data system 18, the majordifferences in the implementation being in the type and quantity of datasent via the external interface circuitry 36, and the needed processingpower at the location where processing is done.

Digital sampling circuitry 260 and spectrum analysis circuitry 262 arecontrolled by timing and control circuitry 22 in a manner similar to thePLL embodiment depicted in FIG. 8. The block diagram in FIG. 15 depictsan alternative embodiment of timing and control circuitry 22, adaptedfor controlling the alternative reader 10 circuitry shown in FIG. 14.PLL timer 48 in FIG. 8 is replaced by Digital Sampling timer 274 in FIG.15. This timer establishes proper sequence and period to the powercontrol 270 and sample start 272 signals to sequence the digitalsampling circuitry 260. The power control signal 270 controls the powerstatus and sleep status of the digital sampling circuitry 260. Thesample start signal 272 causes the digital sampling circuitry 260 togather an appropriate number of samples in a burst sample mode forsending to the spectrum analysis circuitry 262.

Likewise, frequency counter timer 50 in FIG. 8 is replaced by spectrumanalysis timer 280 in FIG. 15. The spectrum analysis timer 280establishes proper sequence and timing to the power control 276 andanalysis start 278 signals, to sequence the spectrum analysis circuitry262. The power control signal 276 controls the power status and sleepstatus of the spectrum analysis circuitry 262. The analysis start signal278 controls the time that the spectrum analysis circuitry 262 beginsevaluating the sample burst 264 provided by the digital samplingcircuitry 260.

Receive circuitry 28 in the alternative embodiment of FIG. 14 isfunctionally and architecturally equivalent to receive circuitry 28 inthe PLL-based embodiment of FIGS. 7 and 10, the only difference beingthe output signal from amplifier stages 86 feeds analog-to-digitalconverter 290 at the input of digital sampling circuitry 260, ratherthan PLL 30.

FIG. 16 is a block diagram depicting an embodiment of digital samplingcircuitry 260. The RF signal from the receive circuitry 28 amplifierstages 86 feeds the input to the analog to digital converter (ADC) 290of the digital sampling circuit 260. The ADC 290 converts the RF signalinto a set of time-related samples taken at close enough intervals andwith sufficient sample quantity to allow the spectrum analysis circuitry262 to achieve its required frequency accuracy. This set of time-relatedsamples will be referred to here as a digital sample burst 264.

The digital sample burst 264 output from the ADC 290 is fed into thespectrum analysis circuitry 262 time to frequency domain conversioncircuit 94, shown in FIG. 17. The internal workings of the frequencydomain conversion 94 are not here specified as this conversion may beany of several means which might include fast or discrete Fouriertransform, discrete or continuous wavelet transform, any of the severalLaplace transforms, any of the several Z-transforms, or other conversionalgorithms known in the art. The internal workings of the frequencydomain conversion 94 may be implemented in hardware or software or anycombination of both to achieve the desired conversion. Since the outputof the frequency domain conversion 94 will be generated at the samplinginterval, and may contain multiple values for transfer to the externaldata interface 17, a result buffer 96 is shown in the spectrum analysiscircuitry 262 to hold these values until they can be transferred to theexternal data interface 17.

In this digital spectrum analysis embodiment, the reader 10 operatingsequence is similar to that described in “Reader Operational Sequence”above, except that the digital sampling circuitry 260 and spectrumanalysis circuitry 262 perform the functions related to determination ofthe ring signal 16 frequency. When the antenna 26 begins to receive thering signal 16, the digital sampling circuit 260 rapidly samples for apredetermined or computed period to obtain a digital sample burst 264.After completion of the digital sample burst 264, the receive circuitry28 and digital sampling circuit 260 are powered down or placed in sleepmode as appropriate. The spectrum analysis circuit 262 converts thedigital sample burst 264 data to frequency domain and places the resultinto the result buffer 96, then is shifted to a low power mode.Subsequently, the timing and control circuitry 22 components are powereddown or placed in sleep mode as appropriate. If programmed for intervalsampling, the timing and control circuitry 22 wakeup timer 38 countsuntil the next sample is due. Otherwise, the timing and controlcircuitry 22 awaits a wakeup command with any other needed instructionsfrom the external interface circuitry 36. The sample data in the resultbuffer 96 is kept available to the external interface circuitry 36 fortransfer to the remote data system 18 as controlled by thecommunications interface.

It will be obvious to anyone skilled in the art that numerous minormodifications may be made to the described digital spectral analysisembodiment to achieve functionally equivalent results. One suchmodification is the use of zero-padding of the ADC data, as is commonpractice with time domain to frequency domain conversions where signalburst data is evaluated. Another such modification is moving thephysical location of the spectrum analysis circuit 262 from the reader10 to the remote data system 18, with ADC 90 data transmitted intime-domain form from the reader 10 to the remote data system 18. Yetanother such modification is frequency converting the ring signal 16 atsome point in the reader 10 by frequency multiplication, division, sum,or difference circuitry, changing the ring signal 16 to an intermediatefrequency signal for any of numerous reasons related to frequencyselectivity, bandwidth, sampling time, etc. Yet another suchmodification is the use of digital signal processing techniques tofilter, shape, analyze, compare with other data, or otherwise processand evaluate the frequency domain or the time domain data.

Likewise, those skilled in the art will readily observe thatcombinations of the various frequency determination methods disclosedherein are possible and may be advantageous in different applications.For example, an analog sample and hold circuit may be used incombination with digital spectral analysis, in order to hold the ringsignal 16 long enough to obtain an adequate sample for digitizing.

In another embodiment, a standard RFID tag, of a type known to those inthe art, may be incorporated with sensor 12. Such tag may have aseparate antenna, and operate at a frequency outside Sensor OperationalRange 220. It can be encoded with configuration information on thesensor 12.

The embodiment of the invention has been described above and, obviously,modifications and alternations will occur to others upon reading andunderstanding this specification. The claims as follows are intended toinclude all modifications and alterations insofar as they are within thescope of the claims or the equivalent thereof.

We claim:
 1. A wireless sensor reader comprising: a transmit circuit configured to generate at least one excitation pulse to cause a wireless sensor to emit at least one response signal corresponding to a sensed parameter value; at least one antenna configured to transmit said at least one excitation pulse and receive said at least one response signal; a receive circuit for amplifying said at least one received response signal; a digital sampling circuit for converting a time interval of said at least one amplified received response signal to a digital representation in the time domain; a spectrum analysis circuitry for converting said digital representation of said at least one response signal to a frequency domain representation; and a frequency domain circuitry for processing of said frequency domain representation to determine said wireless sensor sensed parameter value.
 2. The wireless sensor reader of claim 1, wherein said converting of said digital representation of said response signal to a frequency domain representation is performed at least partially by hardware.
 3. The wireless sensor reader of claim 1, wherein said converting of said digital representation of said response signal to a frequency domain representation is performed at least partially by software.
 4. The wireless sensor reader of claim 1 wherein said response signal is a ring signal.
 5. The wireless sensor reader of claim 1 wherein said wireless sensor reader is a handheld device.
 6. The wireless sensor reader of claim 1 further comprising a battery for powering said wireless sensor reader.
 7. A method of reading a wireless sensor comprising: transmitting at least one excitation pulse to a wireless sensor; receiving at least one response signal from said wireless sensor in response to said excitation pulse; amplifying said at least one response signal; sampling a set of values across a time interval of said at least one response signal to obtain multiple time domain samples; converting said time domain samples to the frequency domain to obtain frequency domain information; and evaluating said frequency domain information to obtain the frequency of said response signal.
 8. The method of claim 7, including the additional step of processing said frequency domain information.
 9. The method of claim 7, including the additional step of analyzing said frequency domain information.
 10. The method of claim 7, including the additional step of storing said frequency domain information.
 11. The method of claim 7, including additional steps of: forwarding said time domain samples to a remote processing device; receiving said time domain samples at said remote processing device; and evaluating said time domain samples at said remote processing device to obtain said frequency of said at least one response signal.
 12. The method of claim 7 wherein said response signal is a ring signal.
 13. The method of claim 7 further comprising the step of powering a reader with a battery. 